Phased array transceiver with built-in transmitter linearization feedback

ABSTRACT

Methods and devices for streamlining phase and amplitude calibration and linearization in RF transceiver circuits including a plurality of switchable transmit and receive processing paths is presented. According to one aspect, switchable feedback paths are provided that can selectively feedback a portion of a transmitted RF signal or a test RF signal for use in the calibration. According to another aspect, the switchable feedback paths include combination of switches and couplers to selectively combine feedback from one or more of the switchable feedback paths. According to another aspect, the switchable feedback paths reuse portions of the receive paths of the plurality of switchable transmit and receive processing paths. The switchable feedback paths can be used to provide a combined feedback RF signal based on one or more transmitted RF signals that can be used as a digital pre-distortion feedback for linearization of the one or more transmitted RF signals.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is related to U.S. application Ser. No.16/455,427 entitled “Phased array transceiver with Built-In PhaseInterferometer” filed on even date herewith and incorporated herein byreference in its entirety. The present application may also be relatedto U.S. Pat. No. 9,717,008, entitled “Integrated Circuit CalibrationArchitecture”, issued Jul. 25, 2017, the disclosure of which isincorporated herein by reference in its entirety.

TECHNICAL FIELD

The present teachings relate to RF (radio frequency) circuits. Moreparticularly, the present teachings relate to methods and apparatusesfor streamlining phase and amplitude calibration and linearization in RFtransceiver circuits that can be used in beam forming and/or steeringapplications.

BACKGROUND

The advent of next-generation 5G standard for millimeter-wave cellularcommunication has pushed the design of hybrid multiple-inputmultiple-output (MIMO) transceiver systems for interfacing with antennaarray to provide directional signal transmission or reception. Signalprocessing techniques, such as beam forming or beam steering that canprovide for the necessary spatial filtering, are implemented by acombination of digital and analog circuit blocks in the transceiversystems. Such transceiver systems may be based on time division duplex(TDD) mode of operation where transmission and reception are performedvia a plurality of transmit paths and receive paths that operate at asame frequency but are activated at different time slots.

Performance in beam forming/steering may be dependent on accuracy bywhich the transceiver can generate phase and amplitude information tothe many elements of the antenna array, and/or on accuracy by which thetransceiver can extract phase and amplitude information from the manyelements of the antenna array. Accordingly, calibration of a phase andamplitude adjustment block in the plurality of transmit and receivepaths of the transceiver becomes a requirement, especially in massproduction environments where wider tolerances in lower cost standardcomponents may invariably introduce performance variation.

Some challenges associated with calibration of the transceiver includethe high number of ports from/to which test signals must beread/injected, as well as the millimeter-wave nature of signals ofinterest. Description of some such challenges and implementationexamples can be found, for example, in the above referenced U.S. Pat.No. 9,717,008. Furthermore, it may be desirable to reduce built-in testcircuits related to the calibration, as such test circuits may not onlyincrease a physical size of the transceiver but may also introduceundesired couplings between the different transmit and receive paths ofthe transceiver. Such couplings in turn may affect performance of thebeam forming/steering, including in applications where (patch) elementsof the antenna are fed with (horizontal and vertical) polarized RFsignals.

Furthermore, requirements for higher power output from power amplifiers(PAs) used in transmit paths of transceivers that are used in, forexample, base stations, may require the PAs to operate away from thepower back-off condition (e.g., linear region). Accordingly, any suchPAs may operate in a non-linear region (i.e., saturation) with resultingoutput phase and amplitude distortion that may negatively affectperformance of the PA in exchange for higher output power and efficiencyof the PA. Such trade-off between efficiency and linearity has beenneutralized by current digital pre-distortion (DPD) schemes which mayrealize a linear response of a combined DPD and PA block by cascadingthe PA and its inverse block. As can be taken from the relatedreferences [1] and [2], which are incorporated herein by reference intheir entirety, such DPD schemes may require a separate feedback pathfrom the output of each PA, and therefore from each transmit path, to aninput of a digital signal processor implementing the DPD functionality.It would be clear to a person skilled in the art that such feedbackpaths may cause the same challenges and problems associated with addedcircuits as described above with reference to the calibration of thephase and amplitude adjustment blocks of the transceiver.

Teachings according to the present disclosure aim to simplify complexityof transceivers used in beam forming/steering applications or otherapplications wherein the transceivers couple to a large number ofantenna elements, thereby overcoming some of the challenges and problemsassociated with, for example, calibrating individual transmit and/orreceive paths and/or instrumenting the transceivers for implementationof DPD schemes as described above.

SUMMARY

According to a first aspect of the present disclosure, a transceivercircuit is presented, the transceiver circuit comprising: a) switchabletransmit and receive RF processing paths, each selectively configured tooperate according to one of: a1) a transmit path for adjusting phase andamplitude of an RF signal for transmission through a respective elementof a plurality of elements of an antenna array, and a2) a receive pathfor adjusting phase and amplitude of an RF signal received through therespective element of the antenna array; b) switchable feedback pathsselectively coupled to the switchable transmit and receive RF processingpaths via switchable elements of said feedback paths; and c) one or moreRF combiners coupled to the switchable feedback paths, wherein saidswitchable feedback paths are selectively configured to operateaccording to at least a transmit path linearization mode, and wherein inthe transmit path linearization mode, i) one or more transmit paths ofthe switchable transmit and receive RF processing paths are each coupledto the respective element of the antenna array, and ii) the switchableelements couple a portion of a phase and amplitude adjusted RF signalthrough each of the one or more transmit paths to the one or more RFcombiners to obtain a combined RF signal that is configured to be usedby a controller for linearization of the one or more transmit paths.

According to a second aspect of the present disclosure, a method forlinearizing RF paths of a transceiver circuit used in a time divisionduplex system is presented, the method comprising a transmit pathlinearization mode, wherein during the transmit path linearization mode:a1) switchable elements of the transceiver circuit couple a portion of aphase and amplitude adjusted RF signal through each of one or moretransmit paths of switchable transmit and receive RF processing paths ofthe transceiver system to one or more RF combiners to obtain a combinedRF signal, while other portion of the phase and amplitude adjusted RFsignal is coupled to a respective element of an antenna array, and a2)the combined RF signal is down converted and a down converted signal isprovided to a controller for linearization of the one or more transmitpaths, and wherein the transceiver circuit comprises: b) a plurality ofthe switchable transmit and receive RF processing paths, eachselectively configured to operate according to one of: b1) a transmitpath for adjusting phase and amplitude of an RF signal for transmissionthrough the respective element of a plurality of elements of the antennaarray, and b2) a receive path for adjusting phase and amplitude of an RFsignal received through the respective element of the antenna array.

BRIEF DESCRIPTION OF DRAWINGS

The accompanying drawings, which are incorporated into and constitute apart of this specification, illustrate one or more embodiments of thepresent disclosure and, together with the description of exampleembodiments, serve to explain the principles and implementations of thedisclosure.

FIG. 1A shows a simplified block diagram of a prior art transceiversystem.

FIG. 1B shows a simplified schematic representation of an exemplarycircuit implementation of the prior art transceiver system of FIG. 1A.

FIG. 1C shows details of an RF processing sub-circuit of the prior arttransceiver system of FIG. 1A.

FIG. 2 shows a simplified block diagram of a prior art digitalpre-distortion (DPD) implementation for a single power amplifier.

FIG. 3 shows a simplified schematic representation of an exemplarycircuit implementation of a transceiver system according to anembodiment of the present disclosure comprising a plurality ofswitchable feedback paths coupled to switchable transmit and receiveprocessing paths of the transceiver system.

FIG. 4A shows details according to an embodiment of the presentdisclosure of elements of a switchable feedback path coupled to a poweramplifier (PA) and a low noise amplifier (LNA) of a switchable transmitand receive processing path of the transceiver system.

FIG. 4B shows the switchable transmit and receive processing path ofFIG. 4A configured to provide a portion of a transmitted signal to aswitchable feedback path, the switchable feedback path independent froma receive path of the switchable transmit and receive processing path.

FIG. 4C shows the switchable transmit and receive processing path ofFIG. 4A configured to provide a test signal to a switchable feedbackpath, the switchable feedback path based on a receive path of theswitchable transmit and receive processing path.

FIG. 4D shows two switchable transmit and receive processing pathscoupled according to an embodiment of the present disclosure, oneconfigured according to the configuration of FIG. 4B to feedback aportion of a transmitted signal, and the other configured according tothe configuration of FIG. 4C to receive the portion of the transmittedsignal as a test signal.

FIG. 4E1 and FIG. 4E2 show exemplary embodiments of couplings to unusedpoles of switches used in the transceiver system.

FIG. 5 shows a configuration according to an embodiment of the presentdisclosure of a power combiner for combining feedback from the pluralityof switchable feedback paths.

FIG. 6A shows elements of the switchable feedback paths configured tocouple a combined feedback of the plurality of switchable feedback pathsto an envelope detector.

FIG. 6B shows elements of the switchable feedback paths configured tocouple a combined feedback of the plurality of switchable feedback pathsto a receive data channel of a digital processing block.

FIG. 7A and FIG. 7B show exemplary graphs respectively representing asum of two RF signals of same phase and of opposite phase.

FIG. 7C shows an exemplary graph representing sensitivity of an envelopeof a sum of two RF signals with respect to a phase difference of the twoRF signals.

FIG. 8A shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths forcalibrating phase and amplitude of transmit paths of the transceiversystem.

FIG. 8B shows another embodiment according to the present disclosure ofa configuration of the plurality of switchable feedback paths forcalibrating phase and amplitude of transmit paths of the transceiversystem.

FIG. 8C shows yet another embodiment according to the present disclosureof a configuration of the plurality of switchable feedback paths forcalibrating phase and amplitude of transmit paths of the transceiversystem.

FIG. 9 shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths forcalibrating phase and amplitude of receive paths of the transceiversystem, based on calibrated transmit paths.

FIG. 10A shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths forproviding a feedback RF signal based on a transmit path RF signal thatcan be used as pre-distortion feedback for linearization of the transmitpath.

FIG. 10B shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths forproviding a combined feedback RF signal based on a combination of RFsignals of a plurality of transmit paths, wherein the combined feedbackRF signal can be used as pre-distortion feedback for linearization ofthe plurality of transmit paths.

FIG. 11 is a process chart showing a method for linearizing RF paths ofa transceiver circuit used in a time division duplex system, the methodcomprising a transmit path linearization mode.

DETAILED DESCRIPTION

Throughout this description, embodiments and variations are describedfor the purpose of illustrating uses and implementations of theinventive concept. The illustrative description should be understood aspresenting examples of the inventive concept, rather than as limitingthe scope of the concept as disclosed herein.

The present disclosure describes electrical circuits (circuitalarrangements) in electronics devices (e.g., cell phones, radios) havinga plurality of devices, such as for example, transistors (e.g.,MOSFETs). Persons skilled in the art will appreciate that suchelectrical circuits comprising transistors can be arranged asamplifiers.

The term “amplifier” as used in the present disclosure may to refer toamplifiers comprising stacked transistors configured as amplifiers orother amplifiers known in the art that do not use stacked transistors,and can be used, for example, as power amplifiers (PAs) and/or low noiseamplifiers (LNAs). An amplifier can refer to a device that is configuredto amplify a signal input to the device to produce an output signal ofgreater magnitude than the magnitude of the input signal. Stackedtransistor amplifiers, in particular stacked transistor amplifiersoperating as a cascode configuration, are described for example in USPatent Application No. 7,248,120, issued on Jul. 24, 2007, entitled“Stacked Transistor Method and Apparatus”, the disclosure of which isincorporated herein by reference in its entirety. As used herein, theterm “amplifier” can also be applicable to amplifier modules and/orpower amplifier modules having any number of stages (e.g., pre-driver,driver, final), as known to those skilled in the art. As used herein theterm “low noise amplifier” or “LNA” are intended to refer to anamplifier comprising a degeneration impedance that comprises aninductor. It is possible that the techniques in this invention apply toa common gate input topology as well.

FIG. 1A shows a simplified block diagram of a prior art transceiversystem (100), including a digital processing block (110), ananalog/digital conversion block (120), an IF-RF block (130), an RFprocessing block (140), a Tx/Rx switch block (150), and an antenna array(160). The transceiver system (100) may include a plurality of transmitand receive paths for respective transmission and reception of aplurality of RF signals via the antenna array (160). As described above,the transceiver system (100) may be used in beam forming/steeringapplications where phase and amplitude of the plurality of RF signalsare controlled to generate constructive interference at selected beamangles radiating from the antenna array and destructive interference atother beam angles. As the beam forming/steering may be used in receptionand transmission, the plurality of transmit paths and the plurality ofreceive paths may include phase and amplitude adjustment blocks. Asshown in FIG. 1B later described, such phase and amplitude adjustmentblocks may be bidirectional, in other words, one such block may providea same signal path during reception and transmission. It should be notedthat as used herein, a block is referred to a circuit designed for aspecific functionality, wherein the circuit may be partitioned accordingto any number of schemes known in the art, including one or more of aboard, a module, an integrated circuit, or other schemes known in theart.

With continued reference to FIG. 1A, as it is well known to a personskilled in the art, the digital processing block (110) may operate inthe digital domain to, for example, generate modulated data signals(e.g., 64, 128, 256 QAM or other) for transmission through the antennaarray (160) and to demodulate received data signals from the antennaarray (160). The analog/digital conversion block (120) may translate themodulated data signals from digital to analog for further processing inthe analog domain by blocks (130, 140, 150) of the transceiver (100)prior to transmission via the antenna array (160), and translate thereceived data signals from analog to digital for further processing inthe digital domain by the digital processing block (110). The IF-RFblock (130) may perform up conversion and down conversion between alower intermediate frequency (IF) for processing in the digital domain(e.g., block 110) and a higher RF frequency (e.g., 28 GHz and higher)for processing in the analog domain (e.g., block 140, 150, 160).

With continued reference to FIG. 1A, the RF processing block (140) mayperform amplitude and phase control of the plurality of RF signals fortransmission/reception via the antenna array. As shown in FIG. 1B laterdescribed, transmission and reception of an RF signal for each of theplurality of transmit and receive paths may respectively be provided viaa respective power amplifier (PA) and low noise amplifier (LNA). TheTx/Rx switch block (150) may selectively couple a transmit path or areceive path of the transceiver to an element of the antenna array (160)for respective transmission or reception of an RF signal via the elementof the antenna array (160). In a case where the antenna array (160) isused for beam forming/steering, the Tx/Rx switch block (150) maysimultaneously couple a plurality of transmit paths or a plurality ofreceive paths for respective transmission or reception of a beam.Furthermore, the Tx/Rx switch block (150) may be used in implementationof time division duplex (TDD) mode of operation of the transceiversystem (100) wherein the transmit paths and receive paths are temporallyseparated.

FIG. 1B shows a simplified schematic of an exemplary circuitimplementation of the prior art transceiver system (100) of FIG. 1A,including blocks (120, 130, 140, 150). In the exemplary implementationshown in FIG. 1B, a single dual-polarization (e.g., horizontal andvertical), channel of the digital processing block (110) is used tocommunicate (interface) with sixty-four elements of the antenna array(160). In other words, the single dual-polarization channel is used togenerate data signals (e.g., transmit data (115 a, 115 b)) forprocessing through sixty-four separate pairs of transmit paths of thetransceiver system (100) prior to transmission via sixty-four elementsof the antenna array (160), and to decode data signals received (e.g.,receive data (112 a, 112 b)) via sixty-four elements of the antennaarray (160) and processed through sixty-four separate pairs of receivepaths of the transceiver system (100). Accordingly, as can be clearlyseen in FIG. 1B, symmetrical circuits are provided for processing (e.g.,transmit and receive) of each of the (horizontal and vertical)polarizations, indicated in the figure with elements designated withsuffices a and b in their respective reference designators. Due to thesymmetrical nature of the implementation, the following description isprovided mainly with respect to one of the symmetrical circuits.

With further reference to FIG. 1B, each of the blocks (120), (130),(140) and (150) includes a main functional element to implement aspecific function of the block. For example, the analog/digitalconversion block (120) includes a digital-to-analog conversion (DAC)circuit (125) to receive data signals (115 a) to be transmitted from theblock (110) and an analog-to-digital conversion (ADC) circuit (122 a) tofeed data signals (112 a) received from the antenna array (160) to theblock (110); the IF-RF block (130) includes a dual polarization mixer(135) for up and down conversion of the data signals; the RF processingblock (140), with further details shown in FIG. 1C later described,includes a plurality (e.g., sixty-four per polarization) of (switchable)transmit and receive RF processing paths (145 a) with adjustableamplitude and phase features to implement, for example, beamforming/steering; and the Tx/Rx switch block (150) includes a plurality(e.g., sixty-four per polarization) of (antenna) switches (155 a) toselectively couple one of the transmit and receive RF processing paths(145 a) to sixty-four elements of the antenna array (160).

With continued reference to FIG. 1B, the blocks (120), (130) and (140)may further include signal conditioning elements such as buffers (e.g.,121 a), filters (e.g., 123 a, 133 a) and power combiners/dividers (e.g.,124 a, 134 a, 144 a), as well as switches (e.g., 122 a), whereinelements in FIG. 1B depicted with same shapes indicate like elements. Itshould be noted that such elements and their application in atransceiver as exemplified in FIG. 1B are well known in the art. Forexample, it is well known in the art to include a low pass filter (e.g.,123 a) after a digital-to-analog conversion (e.g., 125 a) to removedigital conversion noise, or before an analog-to-digital conversion(e.g., (122 a) to remove any unwanted higher frequency components thatmay be present on the signal to be digitized. It is also well known inthe art to include a band pass filter (e.g., 133 a) after up conversionor before down conversion to reject an unwanted image or frequencyspectrum. It is also well known in the art to use buffers (e.g., 121 a)to strengthen signals and/or adapt the signals to reduce loss (e.g.,impedance matching) when coupled to other elements (e.g., 122 a, 123 a).

With continued reference to FIG. 1B, because the blocks (120, 130, 140)communicate with a single data channel (115 a, 112 a) of the block(110), power combiners/dividers (e.g., 124 a, 134 a, 144 a) are used todivide the transmit data signal (115 a) into a plurality of RF signalsfor transmission via elements of the antenna array (160) and to combinea plurality of RF signals received via elements of the antenna array(160) to generate the receive data signal (112 a). A well-known in theart power combiner/divider is a Wilkinson power combiner/divider (e.g.,element (344 a) shown in FIG. 5 later described) which may combine aplurality of RF signals into one RF signal or vice versa. In some cases,the power combiners/dividers may be replaced by a combination of one ormore switches, couplers, and series or parallel impedances that incombination combine a plurality of RF signals into one RF signal ordivide one RF signal into a plurality of RF signals. Variousimplementations of such power combiners/dividers are well known in theart and are outside the scope of the present disclosure.

In order to implement temporal selectivity between transmission andreception of data signals through the blocks (120, 130, 140) shown inFIG. 1B, switches (e.g., 122 a) are used to selectively couplebidirectional signal paths that can be shared during transmission andreception to unidirectional transmit or receive signal paths. Forexample, during transmission, switch (122 a) may couple a(unidirectional) signal path from the transmit data signal (115 a) tothe (bidirectional) low pass filter (123 a), and during reception, theswitch (122 a) may couple a (unidirectional) signal path to the receivedata signal (115 a) to the (bidirectional) low pass filter (123 a). Suchbidirectional signal paths may be provided by elements of the blocks(120, 130, 140) that can operate in both transmit and receive modes,such as, for example, the power combiners/dividers (e.g., 123 a, 133 a,144 a), the filters (e.g., 123 a, 133 a), as well as phase and amplitudeadjustment blocks (e.g., 148 a of FIG. 1C) of the switchable transmitand receive RF processing paths (145 a) as described later withreference to FIG. 1C. Furthermore, although not shown in FIG. 1B, suchelements that provide bidirectional signal paths may be adjustable so toindependently optimize performance of the path when operating in each ofthe transmit and receive direction. Adjustment may be provided by way ofcontrol lines under control of a signal aware controller, such as, forexample, the digital signal processing block (110), and may affect oneor more of an amplitude and phase of a signal passing through the path(which can be different for each direction).

It should be noted that the circuit shown in FIG. 1B may not representtransceiver (110) of FIG. 1A in its entirety but rather may representone part of such transceiver that communicates with one channel (e.g.,dual-polarized transmit data (115 a, 115 b) and receive data (112 a, 112b)) of a plurality of channels of the digital signal processing block(110). Accordingly, the transceiver system may include a plurality ofcircuits similar to one depicted in FIG. 1B, each such circuitcommunicating with one channel of the plurality of channels. Forexample, if the circuit of FIG. 1B communicates with sixty-four elementsof the antenna array (160), four such circuits may be used tocommunicate with an antenna array having two hundred fifty-six (256)elements.

It should be noted that the circuit shown in FIG. 1B represents one ofmany possible architectures for providing a communication interfacebetween elements of the antenna array (160) and the singledual-polarized channel of the digital processing block (110). As shownin the exemplary configuration of FIG. 1B, during transmission, a powercombiner/divider (124 a) of the analog/digital conversion block (120)divides a signal into four signals of substantially equal power whichare each fed to one of four sub-circuits (130′, 130″, 130′″, 130″″) ofthe IF-RF block (130), wherein FIG. 1B shows one such sub-circuit(130′). In turn, sub-circuit (130′) processes one of the four dividedsignals and further divides, via a power divider/combiner (134 a), theprocessed divided signal into four signals of substantially equal powerwhich are each fed to one RF processing sub-circuit (140′) of four RFprocessing sub-circuits (140′, 140″, 140′″, 140″″) of the RF processingblock (140). Finally, sub-circuit (140′) processes one of the four(twice-) divided signals and further divides, via a powerdivider/combiner (144 a), the processed divided signal into four signalsof substantially equal power which are each fed to one of fourswitchable RF processing paths (145 a) of the RF processing sub-circuit(140′).

With continued reference to FIG. 1B, According to the exemplary priorart configuration of FIG. 1B, for each of the two polarizations (e.g.,represented by suffices a and b in the reference designators), there aresixty-four (4×4×4) switchable RF processing paths (145 a) equallypartitioned into four RF processing sub-circuits (140′, 140″, 140′,140″″) which are each equally partitioned into four sub-circuits (130′,130″, 130′, 130″″). During reception, signals in the receive paths arecombined via reverse paths provided by the power dividers/combiners (144a, 134 a, 124 a). Such partitioning may be in view of design goals andperformances which may take into account, for example, a desired levelof integration, cost and physical size. For example, the (four)switchable RF processing paths (145 a) may be monolithically integratedand embedded on a same die that forms a module for each of the foursub-circuits (140′, 140″, 140′, 140″″). In turn, the modules (140′,140″, 140′, 140″″) may be arranged on each of the sub-circuit (130′,130″, 130′, 130″″), etc.

A person skilled in the art would clearly understand that by placing thevarious power dividers/combiners (e.g., 124 a, 134 a, 144 a) atdifferent positions in the blocks (120, 130, 140), differentpartitioning of the communication interface between elements of theantenna array (160) and the single dual-polarized channel ((115 a, 115b), (112 a, 112 b)) of the digital processing block (110) can beobtained. For example, by removing all of the power dividers/combiners(124 a, 134 a, 144 a), the single dual-polarized channel ((115 a, 115b), (112 a, 112 b)) may communicate only to a single element of theantenna array (160) through a single switchable RF processing path (145a), and accordingly sixty-four different channels of the digitalprocessing block (110), each channel communicating via a circuit similarto one shown in FIG. 1B (with only 130′, 140′, and a single 145 a) maybe required to provide same functionality as one provided by the circuitshown in FIG. 1B.

FIG. 1C shows details of the RF processing sub-circuit (140′) comprisinga plurality of the switchable transmit and receive RF processing path(145 a, only one showed for clarity). Switches (142 a, 142 a, 149 a, 155a) determine mode of operation between transmission and reception bycoupling an RF signal path to an element (160 mn) of the antenna array(160) through a power amplifier (PA, 146 a) for transmission, or to alow noise amplifier (LNA, 147 a) for reception. As shown in FIG. 1C, theRF processing sub-circuit (140′) is set for transmission mode ofoperation. Accordingly, switches (142 a, 142 a) are set so that an RFsignal provided via the power divider/combiner (134 a) is routed througha buffer (141 a), then is divided into four RF signals of substantiallyequal power by the power divider/combiner (144 a), each fed to aswitchable transmit and receive RF processing path (145 a). The dividedRF signal passes through a phase and amplitude adjustment block (148 a)where a phase and amplitude of the divided RF signal can be adjusted inview, for example, of a beam forming/steering application. It should benoted that although description according to the present disclosure ismade with respect to both phase and amplitude adjustments of the RFsignal processed by the switchable transmit and receive RF processingpaths (145 a), according to some embodiments, phase adjustment only maybe sufficient to provide desired beam forming/steering performance orother sought after performance. In other words, the block (148 a) maycomprise at least phase adjustment capability. Switches (149 a, 155 a)are set so that the RF signal is amplified by the PA (146 a) andtransmitted to the element (160 mn) of the antenna array (160). Duringreception mode of operation, the switches (142 a, 142 a, 149 a, 155 a)are set to provide a conduction path from the element (160 mn) of theantenna array (160) through the LNA (147 a), the phase and amplitudeadjustment block (148 a), the power divider/combiner (144 a), the buffer(141 a, lower one shown in the figure), and to the powerdivider/combiner (134 a), where the power dividers/combiners (144 a, 134a) respectively combine RF signals from other elements (e.g., 160 kp) ofthe antenna array (160).

FIG. 2 shows a simplified block diagram of a prior art digitalpre-distortion (DPD) implementation for a single power amplifier (246),wherein DPD functionality is implemented via a feedback path (225) thatoperates on a sampled RF signal (250′) output by the amplifier (246) anda digital processing block (220). As can be seen in FIG. 2, a coupler(250) may be used to provide the sampled RF signal (250′) to thefeedback path (225) and fed to an analog to digital block prior toprocessing by the digital processing block (220). In turn, the digitalprocessing block (220) processes the digitized sampled RF signal (250″)according to well known in the art DPD algorithms to generate apre-distortion component that when added to an input signal (215 a)generates a pre-distorted signal (215 a′). When the pre-distorted signal(215 a′) is processed by the analog RF path that includes a mixer (235)for up conversion and the (non-linear) amplifier (246), thepre-distortion component of the pre-distorted signal (215 a′)complements the non-linear response of the amplifier (246) for aneffective output RF signal of the amplifier (246) that is substantiallyfree of non-linearities. In other words, the pre-distortion compensatesfor non-linearities of the amplifier (246).

With further reference to the prior art DPD implementation shown in FIG.2, it would be clear to a person skilled in the art that suchimplementation may be extended to the prior art transceiver system shownin FIG. 1B by replicating the feedback path (225) and functionality ofthe digital processing block (220) for each amplifier (e.g., 146 a ofFIG. 1C) of a plurality of switchable transmit and receive paths of thetransceiver. Some description of such implementation is provided in therelated reference [2] which is incorporated herein by reference in itsentirety. On the other hand, the related reference [1] which isincorporated herein by reference in its entirety, describes animplementation wherein the (single-input single-output, SISO) DPD modelshown in FIG. 2 can be used as a SISO DPD model that operates on acombined over-the-air signal emitted from an antenna array (e.g., 160)that is coupled to the transceiver. However, a person skilled in the artwould clearly understand that such implementations described in therelated references [1] and [2] are impractical for a production system,due for example to the large amount of extra feedback paths required inone implementation and the over the air detector and related positioningmechanism required in the other. As will be later described, teachingsaccording to the present disclosure allow implementation of a simpleSISO DPD model (e.g. per FIG. 2) without the need of an over-the-airdetector and without the need of (complicated) extra feedback paths.

FIG. 3 shows a simplified schematic representation of an exemplarycircuit implementation of a transceiver system (300) according to anembodiment of the present disclosure comprising a plurality ofswitchable feedback paths (e.g., comprising elements 324 a, 332 a, 334a, 344 a, 375 a) coupled to switchable transmit and receive RFprocessing paths (145 a, 145 b) of the transceiver system (300). Aperson skilled in the art would clearly recognize that the transceiversystem (300) includes basic elements of the transmit and receive RFprocessing paths (145 a, 145 b) of the exemplary prior art transceiversystem described above with reference to FIGS. 1A, 1B and 1C to whichelements (324 a, 332 a, 334 a, 375 a) for implementation of theswitchable feedback paths are added. In turn, such switchable feedbackpaths are selectively coupled to an output of an amplifier (e.g., poweramplifier) or input of an amplifier (e.g., LNA) of each of theswitchable transmit and receive RF processing paths (145 a, 145 b). Itshould be noted that such switchable feedback paths according to thepresent teachings may be implemented in any prior art transceiver systemconfiguration operating according to a TDD mode of operation beyond theexemplary configuration of FIG. 1A.

With continued reference to FIG. 3, the switchable feedback paths mayinclude a combination of one or more switches or switching elements(e.g., 332 a, 334 a, 375 a), filters (e.g., within 334 a), and powercombiners (e.g., 324 a, 344 a, and within 334 a). As used herein, aswitching element may refer to an element comprising at least one switchthat is configured to selectively provide at least twodifferent/separate conduction paths. In combination with the basicelements of the transmit and receive processing paths (145 a, 145 b),such switches, filters and power combiners define the switchablefeedback paths according to the present disclosure that may be used toselectively feed information about a phase and amplitude of a signalbeing processed in a receive or transmit path of the transceiver system(300) for further processing. The switchable feedback paths according tothe present teachings may use parts/segments of the transmit and receiveprocessing paths (145 a, 145 b). In other words, the switchable feedbackpaths may share segments of conduction paths of the transmit and receiveprocessing paths (145 a, 145 b).

With continued reference to FIG. 3, according to some exemplaryembodiments, sharing of the conduction paths of a transmit and receiveprocessing path (145 a, 145 b) may be performed while said processingpath is active, such as, for example, sharing of a segment of a transmitpath and/or a receive path during an active transmit phase of theprocessing path. According to further exemplary embodiments, sharing ofthe conduction paths may be performed while the transmit and receiveprocessing path (145 a, 145 b) is not active, such as, for example,sharing (e.g., reusing) of a segment of a transmit path and/or a receivepath of said processing path for a task other than transmit or receivevia the antenna array. As used herein, the term “active” with referenceto a transmit and receive processing path (145 a, 145 b) may refer to acondition wherein the said processing path receives and/or transmits anRF signal via elements of the antenna array. Accordingly, the transmitand receive processing path (145 a, 145 b) may be active during anactive transmit phase wherein an RF signal is transmitted via an elementof the antenna array, and/or during an active receive phase wherein anRF signal is received via an element of the antenna array.

With continued reference to FIG. 3, according to some exemplaryembodiments of the present disclosure, information through the feedbackpaths may be provided to a signal aware controller, such as, forexample, to the digital processing block (110) which in turn can use theinformation for various tasks. According to some exemplary embodimentsof the present disclosure, the various tasks may include phase and/oramplitude calibration of one or more transmit paths of the transceiversystem (300), phase and/or amplitude calibration of one or more receivepaths of the transceiver system (300), and/or linearization of one ormore of the transmit paths of the transceiver system (300).

With continued reference to FIG. 3, according to some exemplaryembodiments of the present disclosure, phase and amplitude calibrationof a transmit or receive path may be based on an RF signal provided bythe switchable feedback path and adjusted via the phase and amplitudeadjustment blocks (e.g., 148 a of FIG. 1C) of the (switchable) transmitand receive RF processing path (e.g., 145 a). According to someexemplary embodiments of the present disclosure, the phase and amplitudeadjustment block (e.g., 148 a) of the transmit and receive RF processingpath (e.g., 145 a) may be adjusted for a transmit path and a receivepath of said processing path independently from one another.

With continued reference to FIG. 3, according to some exemplaryembodiment of the present disclosure, the phase and amplitudecalibration may be based on equalization of phase and amplitude acrossall the transmit and receive processing paths (145 a, 145 b). In otherwords, the processing paths may be calibrated (via settings of theblocks 148 a, 148 b) to reduce phase and amplitude imbalances among thepaths so to provide a substantially same amplitude and phase responsiveto a same RF signal conducted through the paths.

According to an embodiment of the present disclosure, an interferometricprocedure may be used for phase and amplitude calibration of the pathswherein RF signals through the paths may be combined (e.g., via powercombiners) and phase/amplitude adjustments may be performed in view ofdetected/monitored minima/maxima of a combined RF signal. In particular,as later described with reference to FIGS. 7A-7C, phase adjustment maybe performed to obtain a null of an amplitude of the combined RF signal,and amplitude adjustment may be performed to obtain a minimum value ofthe null. According to further exemplary embodiments of the presentdisclosure, as later described with reference to FIGS. 8A-8C, theminima/maxima of the combined RF signal may be detected/monitored via anenvelope detection circuit (e.g., 365 a) which is known per se, or viaany other suitable analog/digital circuit.

With continued reference to FIG. 3, according to some exemplaryembodiments of the present disclosure, the phase and amplitudecalibration may include cross calibration between the two differentpolarizations as later described with reference to FIG. 9. For example,a calibrated transmit path of one polarization can be used to calibratethe receive paths of the other polarization. Such cross calibration maybe possible due to a degree of isolation inherently required betweenconduction paths of the two polarizations of the transceiver system(300), which in turn allows for a reduced cross coupling of an RF signalthrough a transmit path of one polarization that may be used tocalibrate a receive path of the other polarization.

With continued reference to FIG. 3, according to some exemplaryembodiments of the present disclosure, the phase and amplitudecalibration may be performed via small signal amplitude condition of theRF paths, in other words, in view of conducted RF signals that have alow amplitude so to not saturate amplifiers in the RF paths (which cangenerate nonlinearities that may render calibration inaccurate). Smallsignal amplitudes may be provided via test signals generatedspecifically for the calibration during a non-active phase of the RFpaths. Alternatively, or additionally, calibration may be performedduring an active transmit phase in view of portions of a transmitted RFsignal with a small signal amplitude, such as, for example, during apreamble and/or various synchronization frames. A signal awarecontroller, such as the digital processing block (110), may controlstart/stop of the calibration.

With continued reference to FIG. 3, according to some exemplaryembodiments of the present disclosure, as later described with referenceto FIGS. 10A-10B, linearization of one or more transmit paths of thetransceiver system (300) may be based on a combined RF signal from theone or more (active) transmit paths that is fed back, via the switchablefeedback paths, as a receive data signal (e.g., 112 a, 112 b) to thedigital processing block (110). Accordingly, (single-input,single-output, SISO) DPD algorithms embedded within, or availablethrough, the digital processing block (110), may derive pre-distortioncomponents to generate a pre-distorted transmit data signal (e.g., 115a, 115 b). A person skilled in the art would appreciate flexibility andcompactness of such approach for linearizing either a single transmitpath or a plurality of transmit paths using a known in the art SISOtechniques. As linearity may typically change according to a responsetime that is much slower compared to a frame rate of transmitted data,linearization may be performed in a piecewise fashion, wherein differentcombined RF signals corresponding to different distinct groups of thetransmit paths are each linearized at different time slices. Accordingto some embodiments of the present disclosure, such groups maycorrespond to groups of one or more individual columns or rows ofelements of the antenna array (160) used for steering of elevationand/or azimuth of an emitted signal.

With continued reference to FIG. 3, a switchable feedback path accordingto the present disclosure may include a power combiner (344 a) that isselectively coupled, via a switching element (375 a) of the switchablefeedback path, to transmit paths of the (switchable) transmit andreceive RF processing paths (145 a). In the exemplary configurationshown in FIG. 3, the power combiner (344 a) can be selectively coupledto each of the four transmit and receive RF processing paths (145 a) ofthe RF processing sub-circuit (140′). As described earlier and shown inFIG. 3, similar feedback paths are provided for each of the twopolarizations, although description in the present disclosure is mainlyprovided with reference to one of the polarizations. Furthermore, itshould be noted that the switchable feedback path according to thepresent disclosure extends across all the transmit and receive paths ofthe transceiver system configuration shown in FIG. 3. In other words,with exemplary reference to elements (375 a) and (344 a), similarelements (375 a) and (344 a) and respective connections are providedwith respect to each of the processing paths (145 a) and each of the RFprocessing sub-circuits (140′, 140″, 140′, 140″″).

With continued reference to FIG. 3, the switchable feedback pathaccording to the present disclosure may further include a switchingelement (334 a), with further details shown in FIGS. 6A-6B, whichincludes a power combiner (e.g., 334 a 2 of FIG. 6A) that is coupled tothe power combiner(s) (344 a) and a switch (e.g., 334 a 1 of FIG. 6A) toselectively route a combined RF signal to the dual polarization mixer(135) through a (bandpass) filter (e.g., 334 a 3 of FIG. 6A) or toanother power combiner (324 a) that is coupled to the envelope detectioncircuit (365 a). Accordingly, the switching element (334 a) allowsselective feedback of a combined RF signal through one or more transmitpaths to be down converted via the dual polarization mixer (135) andused as a receive data signal (112 a) by the digital processing block(110), or to be potentially further combined with RF signals of othertransmit paths (e.g., contained in 130″, 130′, 130″″) prior to being fedto the envelope detection circuit (365 a). It should be noted that ascan be clearly seen in FIG. 3, with further details in FIGS. 6A-6B, theswitchable feedback path according to the present disclosure may furtherinclude a switching element (332 a) which may selectively couple aninput of the dual polarization mixer (135) to element (133 a) for normaloperation (e.g., reception from antenna array 160) or to the switchingelement (334 a) for operation according to a feedback mode. Furthermore,it should be noted that, as is well known in the art, the dualpolarization mixer (135) may be formed by two separate mixers forrespective up conversion and down conversion, wherein the mixers mayshare a same oscillator (e.g., local oscillator). Furthermore, each ofthe two separate mixers may further include two mixers, one for each ofthe two polarizations.

With reference back to the switching element (375 a), as can be seen inFIG. 3, such element, in combination with its dual switching element(375 b) of the other polarization, may be used to selectively couple atransmit or receive path of a transmit and receive processing path(e.g., 145 a) of one polarization to a receive or transmit path of atransmit and receive processing path (e.g., 145 b) of the otherpolarization. As described later with respect to FIG. 9, suchconfiguration can allow cross calibration between the two differentpolarizations, such as, for example, using of a calibrated transmit pathof one polarization to calibrate a receive path of the otherpolarization. As used herein, a dual element of an element of a transmitand receive path of a first polarization that is coupled to an element(e.g., 160 mn, 160 kp of FIG. 1C) of the antenna array (160) refers toan element of a same function that is part of a transmit and receivepath of the other polarization that is coupled to a same element of theantenna array (160). Accordingly, it would be clear to a person skilledin the art that any element part of one transmit and receive path of thetransceiver shown in FIG. 3 may include a dual element.

With continued reference to FIG. 3, switching elements (332 a, 334 a,375 a, 375 b) in combination with basic switching elements (e.g., 149 a,155 a, 122 a of FIG. 1C) of the transceiver system (300) may be used toconfigure the switchable feedback paths. In other words, a conductionpath provided by a switchable feedback path according to the presentdisclosure may be based on a configuration of the switching elements(332 a, 334 a, 375 a, 375 b) in combination with basic switchingelements (e.g., 149 a, 155 a, 122 a of FIG. 1C) of the transceiversystem (300). As described later, such switching elements may configurethe transceiver system (300) in its entirety, or partially based onfewer than the totality of transmit/receive paths, to operate accordingto at least four modes of operation, including: a) normal transmit orreceive, b) transmit paths phase and amplitude calibration, c) receivepaths phase and amplitude calibration, and d) transmit with DPD feedbackfor linearization. According to some embodiments of the presentdisclosure, some such modes of operation may coexist. For example, a)and b) may coexist during a transmit phase, a) and c) may coexist duringa transmit phase, and a) and d) may coexist during a transmit phase.

FIG. 4A shows details according to an embodiment of the presentdisclosure of elements (375 a 1, 375 a 2, 375 a 3, 375 a 4) comprised inthe element (375 a) of the switchable transmit and receive processingpath (145 a) of FIG. 3. As can be seen in FIG. 4A, such elements includea power coupler (375 a 1) coupled to an output of the amplifier (PA, 146a) and configured to couple a small, such as for example in a range of−20 dB to −30 dB) portion of a power of an RF signal, RF2 a, output bythe amplifier (146 a) to a common (pole) terminal of a switch (375 a 2).Similarly, a power combiner (375 a 3) coupled to an input of theamplifier (LNA, 147 a) is configured to selectively couple and decouplethe input of the amplifier (147 a) to/from an RF signal, RF_(RXin), viaa switch (375 a 4), and to/from an RF signal, RF2 a, received throughthe switch (155 a). It should be noted that positions of switches (149a, 155 a, 375 a 2, 375 a 4) as depicted in FIG. 4A correspond tooperation of the switchable transmit and receive processing path (145 a)during a (normal) transmit phase, wherein switch (149 a) conducts theincoming RF1 a signal that is phase and amplitude adjusted via (148 a)to the amplifier (146 a), and the switch (155 a) conducts an amplifiedversion, RF2 a, of the incoming phase and amplitude adjusted signal fortransmission via an element of the antenna array (160).

With continued reference to FIG. 4A, a person skilled in the art wouldclearly understand that in some cases, it may be desirable to terminatesignal paths that are left opened by unused throws of the switches(e.g., 375 a 2, 375 a 4, 155 a). For example, in the shown position inFIG. 4A, the switch (375 a 2) presents an open circuit to the powercoupler (375 a 1) that is coupled to the output of the amplifier (146a), and the switch (155 a) presents an open circuit to the power coupler(375 a 3) that is coupled to the input of the amplifier (LNA, 147 a).Such open circuits may in turn cause undesired signal reflection in theswitchable transmit and receive processing path (145 a) during operationof the path (e.g., per anyone of the configurations shown in the variousfigures of the present application). It follows that according to someexemplary embodiments of the present disclosure, the open/unused throws(associated to poles/switching terminals of the switches as shown inFIGS. 4E1-4E2) of the switches may be terminated via proper terminationimpedances (e.g., resistors), or even terminated via circuits coupled topoles (e.g., switching terminals) of the switches that may be used, forexample, to monitor power through the path (145 a) during operation intransmit or receive modes. This is shown in FIG. 4E1 and FIG. 4E2,respectively corresponding to the switches (375 a 2, 375 a 4) and (155a), wherein the unused poles (i.e., switching terminals that selectivelyconnect to the common pole/terminal of the switch) of the switches (P0)are shown coupled to respective impedances (Z, Z1, Z2) that may becoupled via respective nodes (N, N1, N2) to a reference ground (fortermination) or to a power monitoring circuit (e.g., a power detector).

FIG. 4B shows the switchable transmit and receive processing path (145a) of FIG. 4A configured to provide a portion (e.g., in a range of −20dB to −30 dB) of the transmitted signal (RF2 a) to a switchable feedbackpath, the switchable feedback path independent from a receive path ofthe switchable transmit and receive processing path (145 a). As can beseen in FIG. 4B, the feedback path is created by closing the switch (375a 2) so to couple the portion of the RF2 a signal as a feedback RFsignal, RF_(FB), to the feedback path. It should be noted that thesignal RF_(FB) is the same as signal RF_(RXOUT) since they both comefrom the same node of the switch (375 a 2). They are given differentnames to indicate their intended routing. As can be taken from FIG. 4B,the feedback path may be established (i.e., switched in) while the(output) switch (155 a) is configured for transmission. It should benoted that the feedback path shown in FIG. 4B may also be establishedwhile the switch (155 a) decouples the output of the amplifier (146 a)from the antenna array in which case the incoming RF1 a signal may be,for example, a calibration test signal.

FIG. 4C shows the switchable transmit and receive processing path (145a) of FIG. 4A configured to provide a test signal, RF_(RXin), to aswitchable feedback path, the switchable feedback path based on areceive path of the switchable transmit and receive processing path (145a). As can be seen in FIG. 4C, the feedback path is created by closingthe switch (375 a 4) so to couple the signal RF_(RXin) to the input ofthe LNA (147 a). It should be noted that the switch (155 a) as shown inFIG. 4C presents an open (i.e. high impedance) with respect to the inputof the LNA (147 a) and therefore may have an adverse effect on the inputof the LNA (147 a) as described above with reference to FIG. 4A, whichmay be overcome by using the switch configuration described above withreference to FIG. 4E2. As can be seen in FIG. 4C, switch (149 a) isconfigured to conduct an amplified version of the RF_(RXin) signal forphase and amplitude adjustment through the phase and amplitudeadjustment block (148 a), which can be processed further down thereceive path shown in FIG. 3. It should be noted that the feedback pathshown in FIG. 4C may be established while the switch (155 a) couples theoutput of the amplifier (146 a) to the antenna array as shown in FIG.4A, or while the switch (155 a) decouples the output of the amplifier(146 a) from the antenna ray. In other words, position of the switch(155 a) with respect to the output of the amplifier (146 a) may notaffect operation of the feedback path shown in FIG. 4C. This is due tothe position of the switch (149 a) which decouples the amplifier (146 a)from a shared transmit and receive segment of the switchable transmitand receive processing path (145 a) that includes element (148 a).

FIG. 4D shows two switchable transmit and receive processing paths (145a, 145 b) coupled according to an embodiment of the present disclosure,wherein the processing path (145 b) is configured according to theconfiguration of FIG. 4B to feedback a portion of a transmitted signal,RF_(RXout), and the other processing path (145 a) is configuredaccording to the configuration of FIG. 4C to receive the portion of thetransmitted signal RF_(RXout) as a test signal. As later described withreference to FIG. 9, such configuration can be used to, for example,calibrate a receive path through the processing path (145 a) via acalibrated transmit path through the processing path (145 b).

FIG. 5 shows a configuration according to an embodiment of the presentdisclosure of the power combiner/divider (344 a), also referred toherein as RF combiner, RF divider, power combiner or power divider,shown in FIG. 3 that is used to combine feedback from a plurality ofswitchable feedback paths coupled to a plurality of switchable transmitand receive processing paths (145 a). According to an exemplaryembodiment of the present disclosure, the power combiner/divider (344 a)may include a plurality of quarter wavelength transmission lines (344 a1, 344 a 2) that are combined according to a tree structure as shown inthe figure. A well known in the art power combiner/divider is of aWilkinson type, description of which is outside the scope of the presentdisclosure.

As can be seen in FIG. 5, feedback paths from each of the switchabletransmit and receive processing paths (145 a) are coupled to respectivenodes (RF_(DIV1), RF_(DIV4)) of the power combiner/divider (344 a) toproduce a combined RF signal (based on switched in/out state of thefeedback paths) at node RF_(COM) of the power combiner/divider (344 a).In the exemplary configuration depicted in FIG. 5, switchable feedbackpaths coupled to nodes (RF_(DIV1), RF_(DIV2)) are switched in to providea feedback RF signals (RF_(FB1), RF_(FB2)), whereas switchable feedbackpaths coupled to nodes (RF_(DIV3), RF_(DIV4)) are switched out andtherefore do not contribute to the combined RF signal at the nodeRF_(COM). As previously noted, a power combiner may be provided by othermeans that one showed in FIG. 5, including, for example, via acombination of one or more switches, couplers and series or parallelimpedances according to well known in the art design techniques.

FIG. 6A shows elements of the switchable feedback paths of FIG. 3configured to couple the combined feedback of the plurality ofswitchable feedback paths (e.g., per FIG. 5) to the envelope detector(365 a). As can be seen in FIG. 6A, the switching element (334 a)includes a switch (334 a 1) which can selectively enable a feedback pathfrom a combination of elements (344 a) to a) the envelope detector (365a) through the power combiner (324 a), or b) to the dual polarizationmixer (135) through the bandpass filter (334 a 3). FIG. 6A shows theswitch (334 a 1) configured for enabling the feedback back to theenvelope detector (365 a), and FIG. 6B shows the switch (334 a 1)configured for enabling the feedback back to the dual polarization mixer(135).

With continued reference to FIG. 6A, according to an embodiment of thepresent disclosure, the switching element (332 a) comprises a switch(332 a 1) which can selectively enable coupling of an inputamplifier/buffer (332 a 2) of the dual polarization mixer (135) to thefeedback path (through switching element (334 a)) or to the switch (132a) as part of normal operation of the transceiver (e.g., switchablefeedback path not enabled). In other words, as can be clearly taken fromFIG. 6A, the switch (332 a 1) allows bypassing a configuration of thedual polarization mixer (135) for normal processing of a transmit orreceive RF signal, to instead receive a (combined) feedback RF signalthrough the switching element (334 a). Because the feedback RF signalmay have an amplitude (i.e., power) greater than a receive RF signalduring normal operation, according to an embodiment of the presentdisclosure the input amplifier/buffer (332 a 2) may comprise a variablegain/attenuation so to not overload a processing circuit of the dualpolarization mixer (135) when coupled to the switchable feedback path.

FIG. 7A and FIG. 7B show exemplary graphs respectively representing asum of two RF signals of same phases and of opposite phases. A personskilled in the art would clearly realize that as shown in the figures,considering two RF signals, RF1 and RF2 having a same frequency content,a sum of the two RF signals is at a maximum when the two RF signals arein phase (i.e., phase difference equal to +/−k*360 degrees per FIG. 7A),and the sum is at a minimum (or a null) when the two RF signals are outof phase (i.e., phase difference equal to +/−k*360+180 degrees per FIG.7B). According to some embodiments of the present disclosure, phasecalibration of the transmit paths and of the receive paths of thetransceiver system (300) of FIG. 3 can be provided by nulling the sum ofRF signals through two paths, one of the two paths being considered as areference path and the other as the target path to be calibrated.Nulling can be performed via adjusting of the phase of the RF signalthrough the target path via a corresponding phase and amplitudeadjustment block (e.g., 148 a of FIG. 4A), and monitoring/detection ofthe null can be provided via, for example, feeding of the sum of the twoRF signals to the envelope detection circuit (e.g., 365 a forcalibration of transmit paths per FIGS. 8A-8C later described) or downconverting (e.g., via the dual polarization mixer 135) the sum of thetwo RF signals and feeding a digitized version of it to the digitalprocessing block (110) as a receive data signal (e.g., for calibrationof receive paths per FIG. 9 later described). The sum of the two RFsignals can be provided via combining of the two RF signals as describedabove with reference to, for example, FIGS. 4-6.

FIG. 7C shows an exemplary graph representing sensitivity of an envelopeof a sum of the two RF signals of FIGS. 7A-7B with respect to a phasedifference of the two RF signals. As can be seen in the figure, a slopeof variation of the sum is substantially greater in a region where thetwo RF signals are out of phase (e.g., phase difference of +/−180degrees) when compared to a region where the two RF signals are in phase(e.g., phase difference of 0 degrees). Accordingly, a more accurateposition of a null, and therefore a reference 180 degrees phasedifference, can be obtained as compared to a position of a maximum(i.e., 0 degrees phase difference). Once the null position, andtherefore a corresponding null setting (e.g., control, control word,control bits, etc.) for the phase of the phase and amplitude adjustmentblock (e.g., 148 a) of the path to be calibrated is established,calibration of the path with respect to the phase can be provided byadding 180 degrees to the null setting. It can be further verified thatthe addition of the 180 degrees results in a maximum of the (envelope ofthe) sum as the addition should bring the two signals in phase.

With reference back to FIG. 7A and FIG. 7B, a person skilled in the artwould clearly understand that amplitude value of a detected null may bea function of a difference in amplitudes between the two RF signals, RF1and RF2. In other words, although a null may be obtained via adjustingof the phase of the target signal to be calibrated, a level of the nullmay be dependent of the difference in amplitudes of RF1 and RF2.Accordingly, amplitude calibration of the transmit paths and of thereceive paths of the transceiver system (300) of FIG. 3 can be providedby minimizing a null value detected through calibration of the phase,wherein the minimizing of the null value can be provided by adjustingthe amplitude of the RF signal through the path to be calibrated via acorresponding phase and amplitude adjustment block (e.g., 148 a of FIG.4A). Accordingly, calibration of the phase and amplitude equalizes phaseand amplitude through the target path and the reference path. Values ofthe phase and amplitude adjustment blocks (148 a, 148 b) may thereforebe considered as offset values with respect to a reference value (e.g.,of the reference path). Accordingly offset values (e.g., weights) forequalization between any two paths can be established. For example, ifthe offset (digital) values between a reference path and a first targetpath is (+3, −1) and if the offset value between the reference path anda second target paths is (−1, +2), then offset values between the firstand the second target paths may be deduced as (−4, +3).

It should be noted that although an envelope detection circuit may belikened by a person skilled in the art to a specific design using adiode to rectify an incoming RF signal, teaching according to thepresent disclosure should not be limited to such specific design, asother circuits (combination of digital and/or analog) that can provideinformation on an envelop of the RF signal may be used as the element(365 a). Such other circuits may not only be used as the element (365 a)shown in FIG. 3, but also within the digital processing block (110) forcalibration of the receive paths as described with reference to FIG. 9later described.

FIG. 8A shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths forcalibrating phase and amplitude of transmit paths of the transceiversystem of FIG. 3, wherein the highlighted regions/elements in the figuretrace signal paths enabled by the switchable feedback paths for thecalibration. As can be seen in FIG. 8A, the calibration can be performedduring an active transmit phase of two transmit paths, one of the twotransmit paths (arbitrarily) used as a reference path and the other usedas a target path for calibration. As previously described, calibrationmay be performed via adjustment of the phase and amplitude adjustmentblock (e.g., 148 a of FIGS. 4A-4D) of a (switchable) transmit andreceive RF processing path (145 a) of the target path. In FIG. 8A, anyone of the two transmit and receive processing paths (145 a) may beconsidered part of the target path and the other, part of the referencepath.

With continued reference to FIG. 8A, an RF signal based on a singletransmit data signal (115 a) is provided to the two transmit and receiveprocessing paths (145 a) via corresponding (highlighted) conductionpaths (i.e., same as normal operation). The switchable element (375 a)of the feedback path coupled to each of the two transmit and receiveprocessing paths (145 a) is set according to FIG. 4A as to provide aportion of the transmitted RF signal (e.g., RF_(FB) per FIG. 4A) to thefeedback path for combination (i.e., summation) through the powercombiner (344 a) per FIG. 5. Furthermore, by setting the switchableelement (375 a) coupled to other transmit and receive paths (145 a) fornormal operation (e.g., feedback paths are switched off) withoutcalibration, as shown in FIG. 4A, a sum of only the two RF signalsthrough the reference path and the target path can be provided to theenvelope detection circuit (365 a). Accordingly, calibration of phaseand amplitude of the target path as described with reference to, forexample, FIGS. 7A-7C can be performed. Once calibrated, the target pathcan be changed to a next/different target path until all (or a group of)the paths are calibrated with respect to one (and a same) referencepath.

With continued reference to FIG. 8A, according to some embodiments ofthe present disclosure, a calibration verification process can bedevised wherein calibration is performed with respect to one or moredifferent reference paths, in other words, the reference path isrotated. Such calibration verification process can reduceinconsistencies in calibration due to, for example, a transmit path witha performance (e.g., with respect to a phase and/or amplitude response)that substantially deviates from performance of a majority of thetransmit paths. It should be noted that in the configuration depicted inthe FIG. 8A, the switchable element (334 a) of the switchable feedbackpath is set according to the configuration depicted in FIG. 6A forrouting of the combined feedback RF signal to the envelope detector (365a).

The calibration verification process according to the present teachingsmay highlight (outlier) transmit paths of higher performance deviation.Such higher performance deviation may be due to variation in performanceof corresponding power amplifiers (e.g., 146 a of FIG. 4A). A personskilled in the art would clearly understand that such variation inperformance may be due to a phase and/or amplitude response of the poweramplifiers operating in saturation. It follows that according to anexemplary embodiment of the present disclosure, the power amplifiers ofthe outlier transmit paths may be operated away from their respectivesaturation regions so to bring their performance in line with poweramplifiers of other transmit paths.

Because of a relatively high amplitude of an RF signal in a transmitpath of the transceiver system (300), a higher isolation betweenconduction paths of the two polarizations is desired. Accordingly, thefeedback paths of the present teachings may not couple (e.g., via powercombiners/dividers, switchable elements, etc.) transmit paths ofdifferent polarizations, as such couplings, even if switchable, mayaffect the higher isolation. It follows that according to an embodimentof the present disclosure, phase and amplitude calibration of thetransmit paths as described above may be performed within a respectivepolarization. This is shown in FIG. 8A wherein the suffix a indicatespaths of same polarization. According to further embodiments of thepresent disclosure, calibration of the transmit paths of the twopolarizations may be performed in parallel. In other words, while thetransmit paths of a first polarization are being calibrated, transmitpaths of the second polarization may be calibrated concurrently.

With continued reference to FIG. 8A, calibration of the transmit pathsmay be provided for transmit paths having corresponding (switchable)transmit and receive RF processing paths (145 a) part of a same RFprocessing block (e.g., 140′) as shown in FIG. 8A, or part of differentRF processing blocks (e.g., any one of 140′, 140″, 140′″ and 140″″ ofFIG. 3) as shown in FIG. 8B, or part of different IF-RF blocks (e.g.,any one of 130′, 130″, 130′″ and 130″″ of FIG. 3) as shown in FIG. 8C.In other words, as the exemplary configuration of FIG. 3 can bepartitioned in many different ways (as previously described),calibration of the transmit paths (and receive paths) according to thepresent teachings can be equally applied for any partitioning of theblocks of the transceiver system (300) shown in FIG. 3.

FIG. 9 shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths forcalibrating phase and amplitude of receive paths of the transceiversystem of FIG. 3 based on the calibrated transmit paths, wherein thehighlighted regions/elements in the figure trace signal paths enabled bythe switchable feedback paths for the calibration. Similar to thecalibration of the transmit paths described above with reference toFIGS. 8A-8C, calibration of the receive paths according to the presentteachings may be performed by feeding a (substantially) same RF signalto a reference receive path and to a target receive path, equalizingphase and amplitude of the target receive path to the reference receivepath via a corresponding phase and amplitude adjustment block (e.g., 148a of FIGS. 4A-4D) and repeating the calibration for each of the targetreceive paths using the same reference receive path. Once done, thereference receive path may be rotated to verify the calibration in amanner similar to one described above with reference to the calibrationof the transmit paths. Accordingly, calibration of the phase andamplitude equalizes phase and amplitude through the target paths and thereference path. Values of the phase and amplitude adjustment blocks (148a, 148 b) may therefore be considered as offset values with respect to areference value (e.g., of the reference path). Accordingly, offsetvalues (e.g., weights) for equalization between any two paths can beestablished. It would be clear to a person skilled in the art that suchoffset values relate to equalization of the paths when operating in areceive mode and may be different from the offset values derived foroperation in a transmit mode as described above with reference to forexample FIGS. 8A-8B.

With continued reference to FIG. 9, calibration of the phase andamplitude of the receive paths according to the present teachings may beperformed within a same polarization. For example, as can be seen inFIG. 9, the reference receive path and the target receive path eachincludes a (switchable) transmit and receive RF processing path (145 a)that is part of the same polarization. Each of such transmit and receiveRF processing path (145 a) includes a respective phase and amplitudeadjustment block (148 a), wherein in the figure, one is shown in adotted box. Each transmit and receive RF processing path of thereference and target receive paths (145 a) is configured according tothe configuration of FIG. 4C for receiving a test signal (e.g.,RF_(RXin) of FIG. 4C). Accordingly, calibration of the phase andamplitude of the receive paths can be performed while the receive pathsare not active (e.g. coupled to the antenna array 160).

With continued reference to FIG. 9, calibration of the phase andamplitude of the receive paths according to the present teachings isprovided by reusing the receive paths while not active. Accordingly, ascan be seen in FIG. 9, the sum of the RF signals through the referenceand the target receive paths is conducted through a normal receive pathof the transceiver system (e.g., 300 of FIG. 3) which is modified toinclude the switching element (332 a) described above with reference to,for example, FIGS. 6A-6B. The down converted sum is provided to thedigital signal processing block (110) as a receive data signal whosephase and amplitude can be detected via circuits that are part, or cancommunicate with, the digital signal processing block (110).Calibration, e.g., equalization, of the phase and amplitude of thetarget receive path can be provided in a manner similar to thecalibration of the target transmit path, wherein a null is firstdetected by sweeping the phase of the RF signal through the target path,and then a level of the null is minimized by sweeping thegain/attenuation of the RF signal through the target path.

With further reference to FIG. 9, according to an embodiment of thepresent disclosure the RF signal through the reference and targetreceive paths may be based on RF signals through equalized/calibratedtransmit paths of a polarization (e.g., polarization b per FIG. 9)different from the polarization (e.g., polarization a per FIG. 9) of thereference and target receive paths. This is shown in FIG. 9, wherein thereference and target receive paths each receive a respective RF signalfrom a corresponding (e.g., dual) calibrated transmit path whichincludes a (switchable) transmit and receive RF processing path (145 b).Accordingly, elements (145 a) of the reference and target receive pathsand elements (145 b) of the corresponding transmit paths are configuredand coupled according to the description above with reference to FIG.4D.

As can be seen in FIG. 9, according to an exemplary embodiment of thepresent disclosure, the calibration of the receive paths may beperformed during an active transmit phase (i.e., transmit mode ofoperation) of the two corresponding transmit paths that are used toprovide equalized RF signals to the receive paths. According to anotherexemplary embodiment of the present disclosure, calibration of thereceive paths may be based on RF signals through the correspondingtransmit paths, the RF signals having small signal amplitudes that maybe provided via test signals generated specifically for the calibrationduring a non-active phase of the transmit and receive RF paths.

With continued reference to FIG. 9, it should be clear to a personskilled in the art that the corresponding transmit paths used in thecalibration of the receive paths may be configured to provide to thereference and target receive paths RF signals having a substantiallysame phase and amplitude. This can be accomplished based on thecalibrated phase and amplitude values of the phase and amplitudeadjustment blocks (e.g., 148 b of FIG. 4D) of the corresponding transmitpaths. According to an exemplary embodiment of the present disclosure,the transmit paths are respective dual paths of the receive pathparticipating in the calibration so to conserve/maintain a desiredhigher isolation across the transmit and receive paths of thetransceiver system of FIG. 3 while providing means (feedback paths) forcalibration.

FIG. 10A shows an embodiment according to the present disclosure of aconfiguration of the plurality of switchable feedback paths of thetransceiver system of FIG. 3 for providing a portion of an RF signalthrough an active transmit path as a pre-distortion feedback that can beused for linearization of the transmit path. As can be seen in FIG. 10A,the switchable elements (375 a, 334 a, 332 a) of a feedback path coupledto the (switchable) transmit and receive RF processing path (145 a) ofthe active transmit path are configured according to the configurationsdescribed above with reference to FIG. 4B and FIG. 6B.

With continued reference to FIG. 10A, a feedback RF signal that is basedon (e.g., a portion of) an RF signal that is amplified by the transmitand receive RF processing path (145 a) is fed back, via the switchablefeedback path, as a receive data signal (e.g., 112 a) to the digitalprocessing block (110). As described above with reference, for exampleto FIG. 2 and FIG. 3, such feedback RF signal can be used to linearizethe transmit path using known in the art SISO DPD algorithms. Althoughthe exemplary configuration of FIG. 10A shows a feedback based on asingle active transmit path, teachings according to the presentdisclosure can extend to a configuration wherein the feedback RF signalis based on a combination (i.e., summation) of portions of RF signalsamplified by a plurality of active transmit paths as shown in FIG. 10B.In other words, as previously described, a known in the art SISO DPDalgorithm operating on a combined RF signal from the plurality of activetransmit paths shown in FIG. 10B may be used as a pre-distortionfeedback to linearize the plurality of transmit paths.

With continued reference to FIGS. 10A and 10B, according to anembodiment of the present disclosure, phase and amplitude of the activetransmit paths that provide the feedback RF signal as pre-distortionfeedback may be calibrated according to the phase and amplitudecalibration techniques of the present teachings. Accordingly, thefeedback RF signal that is used as a “model” for characterization of acombined distortion of the plurality (or a group) of active transmitpaths may be referred to as a weighted summation of RF signals througheach of the plurality (or group) of active transmit paths, each with aweight that is equal to corresponding calibration offset values.

A person skilled in the art is well aware of different methods forimplementing beam forming and/or steering, wherein spatial control(e.g., steering angle) of a combined emitted beam may be performed withrespect to an elevation and/or an azimuth of the beam. Such control maybe achieved via phase and amplitude control of a (dual polarized)transmitted RF signal to each of the elements (e.g., 160 mn, 160 mk, . .. of FIG. 1C) of the antenna array (160). In cases wherein the beamforming/steering can maintain a small variation in the elevation orazimuth, the elements of the antenna array (160) may be groupedaccording to a same column or row, each such grouping capable ofemitting concurrently a different beam according to a substantially sameelevation or azimuth. In such cases, the feedback RF signal may be basedon such grouping, such as to separately linearize the groups. In caseswherein the beam forming includes large variations in elevation and/orazimuth, a signal aware controller may group the transmit path accordingto groups with similar or close elevation and/or azimuth for piecewiselinearization.

FIG. 11 is a process chart (1100) showing various steps of a method forcalibrating RF paths of a transceiver circuit used in a time divisionduplex system, the method comprising a transmit path linearization modeaccording to the present teachings. As can be seen in the process chart(1100), during the transmit path linearization mode, the methodcomprises: switchable elements of the transceiver circuit couple aportion of a phase and amplitude adjusted RF signal through each of oneor more transmit paths of switchable transmit and receive RF processingpaths of the transceiver system to one or more RF combiners to obtain acombined RF signal, while other portion of the phase and amplitudeadjusted RF signal is coupled to a respective element of an antennaarray, per step (1110), and the combined RF signal is down converted anda down converted signal is provided to a controller for linearization ofthe one or more transmit paths, per step (1120).

Based on the above description, a person skilled in the art wouldrealize that the transceiver system described above may be used not onlyin beam forming/steering applications, but also in other RF applicationswherein the transceiver is coupled to a plurality of antenna elements.Furthermore, it is noted that the herein described elements andinstrumentation of the disclosed transceiver system may also be used inother applications beyond the described calibration of thetransmit/receive paths and linearization of the transmit paths. Suchother applications may include, for example, built-in system test(BIST), wherein the transceiver system may (automatically) perform uponpower up checking of various functions and performances of the systemand generate corresponding status (flags).

Reduced layout size advantage provided by the configurations accordingto the present teachings may allow further reduction of a monolithicallyintegrated circuit using such configurations. A person skilled in theart would realize that monolithic integration of any of theconfigurations described above, either in their entireties or partially,may be possible as well, depending on desired implementation goals.

Applications that may include the novel apparatus and systems of variousembodiments include electronic circuitry used in high-speed computers,communication and signal processing circuitry, modems, single ormulti-processor modules, single or multiple embedded processors, dataswitches, and application-specific modules, including multilayer,multi-chip modules. Such apparatus and systems may further be includedas sub-components within a variety of electronic systems, such astelevisions, cellular telephones, personal computers (e.g., laptopcomputers, desktop computers, handheld computers, tablet computers,etc.), workstations, radios, video players, audio players (e.g., mp3players), vehicles, medical devices (e.g., heart monitor, blood pressuremonitor, etc.) and others. Some embodiments may include a number ofmethods.

The term “MOSFET”, as used in this disclosure, means any field effecttransistor (FET) with an insulated gate and comprising a metal ormetal-like, insulator, and semiconductor structure. The terms “metal” or“metal-like” include at least one electrically conductive material (suchas aluminum, copper, or other metal, or highly doped polysilicon,graphene, or other electrical conductor), “insulator” includes at leastone insulating material (such as silicon oxide or other dielectricmaterial), and “semiconductor” includes at least one semiconductormaterial.

As should be readily apparent to one of ordinary skill in the art,various embodiments of the invention can be implemented to meet a widevariety of specifications. Unless otherwise noted above, selection ofsuitable component values is a matter of design choice and variousembodiments of the invention may be implemented in any suitable ICtechnology (including but not limited to MOSFET structures), or inhybrid or discrete circuit forms. Integrated circuit embodiments may befabricated using any suitable substrates and processes, including butnot limited to standard bulk silicon, silicon-on-insulator (SOI), andsilicon-on-sapphire (SOS). Unless otherwise noted above, the inventionmay be implemented in other transistor technologies such as bipolar,GaAs HBT, GaN HEMT, GaAs pHEMT, and MESFET technologies. However, theinventive concepts described above are particularly useful with anSOI-based fabrication process (including SOS), and with fabricationprocesses having similar characteristics. Fabrication in CMOS on SOI orSOS enables low power consumption, the ability to withstand high powersignals during operation due to FET stacking, good linearity, and highfrequency operation (i.e., radio frequencies up to and exceeding 50GHz). Monolithic IC implementation is particularly useful sinceparasitic capacitances generally can be kept low (or at a minimum, keptuniform across all units, permitting them to be compensated) by carefuldesign.

Voltage levels may be adjusted or voltage and/or logic signal polaritiesreversed depending on a particular specification and/or implementingtechnology (e.g., NMOS, PMOS, or CMOS, and enhancement mode or depletionmode transistor devices). Component voltage, current, and power handlingcapabilities may be adapted as needed, for example, by adjusting devicesizes, serially “stacking” components (particularly FETs) to withstandgreater voltages, and/or using multiple components in parallel to handlegreater currents. Additional circuit components may be added to enhancethe capabilities of the disclosed circuits and/or to provide additionalfunctional without significantly altering the functionality of thedisclosed circuits.

A number of embodiments according to the present disclosure have beendescribed. It is to be understood that various modifications may be madewithout departing from the spirit and scope of such embodiments. Forexample, some of the steps described above may be order independent, andthus can be performed in an order different from that described.Further, some of the steps described above may be optional. Variousactivities described with respect to the methods identified above can beexecuted in repetitive, serial, or parallel fashion.

It is to be understood that the foregoing description is intended toillustrate and not to limit the scope of the disclosure, which isdefined by the scope of the following claims, and that other embodimentsare within the scope of the claims. (Note that the parenthetical labelsfor claim elements are for ease of referring to such elements, and donot in themselves indicate a particular required ordering or enumerationof elements; further, such labels may be reused in dependent claims asreferences to additional elements without being regarded as starting aconflicting labeling sequence).

LIST OF REFERENCES

-   Reference [1]: “Single-Input Single-Output Digital Predistortion of    Power Amplifier Arrays in Millimeter Wave RF Beamforming    Transmitters” by Eric Ng et al., Department of Electrical and    Computer Engineering, University of Waterloo, Waterloo, ON, Canada,    published in 2018 IEEE/MTT-S International Microwave Symposium.-   Reference [2]: “A Digital Predistortion Scheme Exploiting    Degrees-of-Freedom for Massive MIMO Systems” by Miao Yao et al.,    Dept. of Electrical and Computer Engineering, Virginia Tech,    published as arXiv:1801.06023v1 [eess.SP] 18 Jan. 2018.

The invention claimed is:
 1. A transceiver circuit comprising: a)switchable transmit and receive RF processing paths, each selectivelyconfigured to operate according to one of: a1) a transmit path foradjusting phase and amplitude of an RF signal for transmission through arespective element of a plurality of elements of an antenna array, anda2) a receive path for adjusting phase and amplitude of an RF signalreceived through the respective element of the antenna array; b)switchable feedback paths selectively coupled to the switchable transmitand receive RF processing paths via switchable elements of said feedbackpaths; and c) one or more RF combiners coupled to the switchablefeedback paths, wherein said switchable feedback paths are selectivelyconfigured to operate according to at least a transmit pathlinearization mode, and wherein in the transmit path linearization mode,i) one or more transmit paths of the switchable transmit and receive RFprocessing paths are each coupled to the respective element of theantenna array, and ii) the switchable elements couple a portion of aphase and amplitude adjusted RF signal through each of the one or moretransmit paths to the one or more RF combiners to obtain a combined RFsignal that is configured to be used by a controller for linearizationof the one or more transmit paths.
 2. The transceiver circuit accordingto claim 1, wherein the combined RF signal is used by the controller asa pre-distortion feedback to a single input single output (SISO) digitalpre-distortion algorithm.
 3. The transceiver circuit according to claim2, wherein the transceiver circuit further comprises a mixer circuitthat is configured, during normal operation of the transceiver, to downconvert the combined RF signal received from the plurality of elementsof the antenna array for decoding of the corresponding received data,and wherein in the transmit path linearization mode, i) the switchableelements couple an input of the mixer circuit to the combined RF signalthat is used as the pre-distortion feedback, and ii) the mixer circuitgenerates a down converted signal corresponding to the said combined RFsignal for use by the controller.
 4. The transceiver circuit accordingto claim 3, wherein based on the down converted signal, the SISO digitalpre-distortion algorithm generates pre-distortion components that areused to generate a pre-distorted transmit data signal that is upconverted by the mixer circuit and transmitted through the one or moretransmit paths that are each coupled to the respective element of theantenna array.
 5. The transceiver circuit according to claim 3, whereinthe one or more transmit paths comprises a plurality of transmit paths.6. The transceiver circuit according to claim 5, wherein the pluralityof transmit paths represent a group of transmit paths that form acombined emitted RF beam through a respective group of elements of theantenna array according to a substantially same elevation or azimuth. 7.The transceiver circuit according to claim 6, wherein the respectivegroup of elements of the antenna arrays form a row or a column of theantenna array.
 8. The transceiver circuit according to claim 6, whereindifferent groups of transmit paths that form combined emitted RF beamswith different substantially same elevation or azimuth are eachlinearized through the transmit path linearization mode at a differenttime.
 9. The transceiver circuit according to claim 1, wherein each ofthe one or more transmit paths comprises an amplifier that is configuredto output a phase and amplitude adjusted RF signal, and wherein theswitchable elements comprise: a power coupler that is coupled to anoutput of the amplifier, the power coupler configured to generate fromthe output of the amplifier the portion of the phase and amplitudeadjusted RF signal; and a first switch that is coupled via a common poleof the first switch to a terminal of the power coupler that carries theportion of the phase and amplitude adjusted RF signal, the first switchconfigured to selectively couple the portion of the phase and amplitudeadjusted RF signal to the one or more RF combiners.
 10. The transceivercircuit according to claim 9, wherein each of the switchable transmitand receive paths comprises an antenna switch that is configured toselectively couple one of the transmit path and the receive path to asame element of the plurality elements of the antenna array, and whereinthe antenna switch is coupled to a terminal of the power coupler thatcarries a remaining portion of the phase and amplitude adjusted RFsignal output by the amplifier.
 11. The transceiver circuit according toclaim 1, wherein the switchable transmit and receive RF processing pathsare grouped in pairs according to a first polarization and a secondpolarization, and wherein the transmit path linearization mode fortransmit paths associated to the first polarization is independent fromthe transmit path linearization mode for transmit paths associated tothe second polarization.
 12. The transceiver circuit according to claim11, wherein the transmit path linearization mode for transmit pathsassociated to the first polarization is enabled concurrently with thetransmit path linearization mode for transmit paths associated to thesecond polarization.
 13. The transceiver circuit according to claim 1,wherein transistors of the transceiver circuit comprisemetal-oxide-semiconductor (MOS) field effect transistors (FETs).
 14. Thetransceiver circuit according to claim 13, wherein said transistors arefabricated using one of: a) silicon-on-insulator (SOI) technology, b)silicon-on-sapphire (SOS) technology, and c) bulk silicon (Si)technology.
 15. The transceiver circuit according to claim 1, whereintransceiver circuit is monolithically integrated.
 16. An electronicmodule comprising the transceiver circuit of claim
 1. 17. A radiofrequency (RF) front-end communication system, comprising: a transmitterand receiver section for selectively transmitting and receiving aplurality of RF signals to/from the plurality of elements of the antennaarray, the transmitter and receiver section comprising the transceivercircuit of claim
 1. 18. The radio frequency (RF) front-end communicationsystem according to claim 17, wherein said system is used in a phasedarray.
 19. The radio frequency (RF) front-end communication systemaccording to claim 18, wherein said system processes RF signals togenerate a beam forming or beam steering of the phased array.
 20. Amethod for linearizing RF paths of a transceiver circuit used in a timedivision duplex system, the method comprising a transmit pathlinearization mode, wherein during the transmit path linearization mode:a1) switchable elements of the transceiver circuit couple a portion of aphase and amplitude adjusted RF signal through each of one or moretransmit paths of switchable transmit and receive RF processing paths ofthe transceiver system to one or more RF combiners to obtain a combinedRF signal, while other portion of the phase and amplitude adjusted RFsignal is coupled to a respective element of an antenna array, and a2)the combined RF signal is down converted and a down converted signal isprovided to a controller for linearization of the one or more transmitpaths, and wherein the transceiver circuit comprises: b) a plurality ofthe switchable transmit and receive RF processing paths, eachselectively configured to operate according to one of: b1) a transmitpath for adjusting phase and amplitude of an RF signal for transmissionthrough the respective element of a plurality of elements of the antennaarray, and b2) a receive path for adjusting phase and amplitude of an RFsignal received through the respective element of the antenna array. 21.The method for linearizing RF paths according to claim 20, the methodfurther comprising: use of the down converted signal as a pre-distortionfeedback to a single-input single-output (SISO) digital pre-distortionalgorithm, wherein the SISO digital pre-distortion algorithm generatespre-distortion components that are used to generate a pre-distortedtransmit data signal that is up converted and transmitted through theone or more transmit paths for linearization of the one or more transmitpaths that are each coupled to the respective element of the antennaarray.